In the field of infrared detectors, the use of devices designed in the form of an array which are capable of operating at ambient temperature, i.e. which do not require cooling down to very low temperatures, is known—these are contrasted with detection devices referred to as “quantum infrared detectors” which can only operate at very low temperature, typically at the temperature of liquid nitrogen.
These uncooled detectors traditionally use the variation in a physical unit of an appropriate material as a function of temperature at around 300 K. In the case of bolometric detectors, this physical unit is electrical resistivity.
Such an uncooled detector is generally associated with:                means of absorbing the infrared radiation and converting it into heat;        means of thermally insulating the detector so that its temperature can rise due to the effect of the infrared radiation to be detected;        thermometric means which, in the context of a bolometric detector, use a resistance element;        and means of reading electrical signals provided by the thermometric means.        
Detectors intended for infrared imaging are conventionally produced as a one- or two-dimensional array of elementary detectors, said array being “monolithically” formed or mounted on a substrate generally made of silicon which incorporates means of sequentially addressing the elementary detectors and means of electrically exciting (stimulating) and of pre-processing the electrical signals generated by these elementary detectors. These means of sequential addressing, electrical excitation and pre-processing are formed on the substrate and constitute a read circuit.
In order to obtain a scene using this detector, the scene is projected through suitable optics onto the array of elementary detectors, each of them constituting an image dot or pixel, and clocked electrical stimuli are applied via the read circuit to each of the elementary detectors or to each row of such detectors in order to obtain an electrical signal which is an image of the temperature reached by each of said elementary detectors. This signal is then processed to a greater or lesser extent by the read circuit and then, if applicable, by an electronic device outside the package in order to generate a thermal image of the observed scene.
The essential difficulty encountered when using bolometric detectors is the extremely small relative variation in their electrical resistance which is representative of the local temperature variations in an observed scene relative to the average value of these resistances.
The constructionally dictated presence of a finite thermal resistance between the bolometer and the substrate means that the temperature of the bolometer is influenced by the temperature of the substrate much more sensitively than temperature variations due to the incident flux which are the only variations to be taken into account from the point of view of the signal to be detected. Residual fluctuations in the temperature of the substrate under normal thermal stabilization conditions, all the more so if the detector does not have such a thermal stabilization system as is increasingly the case with this type of detector in order to reduce its cost, consequently produce an unwanted component in the signal obtained from the bolometer which adversely affects the quality of the signal. Conventionally, the substrate is thermally controlled in order to prevent or at least limit this effect.
In addition, “compensation” structures are used in order to minimize the effects of the temperature of the focal plane on the detector's response. These structures, which are usually bolometers referred to as “blind bolometers”, i.e. bolometers which are not sensitive to the incident optical flux but which are sensitive to the temperature of the substrate, are used in order to generate a so-called compensation current which is subtracted from the current obtained from the imaging bolometers, i.e. the detection bolometers, due to the way in which the electronic circuit is configured.
These compensation structures are typically built so that they have a very low thermal resistance relative to the substrate, unlike the imaging bolometers.
This way, most of the current referred to as “common-mode current”, i.e. current which is not representative of information originating from the scene to be detected, is eliminated.
Also and advantageously, because the compensation structures are substantially at the same temperature as the read circuit and therefore the focal plane, this actually allows significant rejection of any fluctuations in the temperature of the focal plane. Arranging these compensation structures “identically” and repetitively in each column of the array so as to reduce the complexity and overall dimensions of the circuit is a known tactic.
Every bolometer column is sequentially compensated by the same compensation structure when the image is electronically scanned one row at a time. However, compensation structures naturally exhibit spatial variations in resistance because of the technology processes used in their fabrication (which normally originate from the semiconductor industry).
In addition, blind bolometers, like imaging bolometers, as well as certain functions of the read circuit, are affected by noise phenomena in general and so-called “l/f” noise in particular. l/f noise typically produces low-frequency drift, especially very low frequency drift, of the output level of the sensors which adversely affects the quality of the imager. The columnar arrangement of the compensation structures has a negative impact on the quality of the image because of low-frequency variations in the compensated signal which are asynchronous from one column to the next. Besides any special design and implementation measures taken in order to reduce this variability, compensation algorithms must, generally speaking, be developed and applied at the output of the imager in order to improve image quality.
Read circuits for resistive bolometric detectors which use blind bolometers are described, for instance, in the following applications:                “Uncooled amorphous silicon technology enhancement for 25 μm pixel pitch achievement>>, E. MOTTIN et al; Infrared Technology and Applications XXVIII, SPIE Vol. 4820;        “320×240 uncooled microbolometer 2D array for radiometric and process control applications” B. FIEQUE et al; Optical Systems Design Conference, SPIE 5251, September 29;        “Low cost amorphous silicon based 160×120 uncooled microbolometer 2D array for high volume applications” C. TROUILLEAU et al; Optical Systems Design Conference SPIE 5251-16.        
The principle of reading an active array of bolometers is shown schematically in relation to FIG. 1.
Pixel 1 (the term “pixel” is construed here, by extension, as denoting all the structures located so that they are under the influence of one elementary detection point) comprises an active bolometer 2, an NMOS charge injection transistor 3 and a switch 4 which connects pixel 1 to read column 5 and is represented here by a dashed line. Compensation structure 6, which is also referred to as a base clipper in the terminology used in the technical field in question, comprises a blind bolometer 7 connected to power supply VSK and PMOS charge injection transistor 8. During normal operation, the PMOS transistor is in saturation mode. Its current Icomp which flows in the compensation arm is defined by the expression:
  Icomp  =      Vcomp    Rcomp  where:                Vcomp denotes the voltage across the terminals of compensation bolometer 7;        Rcomp denotes the resistance of said compensation bolometer.        
The current which flows through the active arm which comprises NMOS charge injection transistor 3 is expressed by the relation.
  Iac  =      Vac    Rac  where:                Iac denotes the current of the active arm;        Vac denotes the voltage across the terminals of active bolometer 2;        Rac denotes the resistance of said active bolometer.        
The bias voltages of the MOS charge injection transistors are chosen so that, in the absence of any incident scene light flux, i.e. for example when the system is optically shuttered, the difference in current dI=Icomp−Iac between the active arm and the blind compensation arm is substantially zero.
Reading an active bolometer is a two-phase operation. The first phase involves actuating “reset” switch 9 which short-circuits integration capacitance 10 of operational amplifier 11. This gives:Vout=VBUS 
Read column 5 shown by dashed line 5 is therefore brought to the potential VBUS. “Reset” switch 9 is then opened and “select” switch 4 is closed to connect pixel 1 to read column 5. Current difference dI is integrated by capacitance Cint 10 over finite integration time Tint. Integration produces an output voltage level referred to as “continuous level” or NC in the reference case where a uniform temperature scene is observed, this typically reveals the variability of the imaging array. This is the standard method for characterizing the reading of active bolometers.
  NC  =      VBus    -                            T          ⁢                                          ⁢          int                          C          ⁢                                          ⁢          int                    ⁢      dI      
Bolometers are biased so as to ensure both a dynamic output signal response and efficient compensation.
A more rigorous expression would be obtained by considering, for the last term, the integral of the function dI(t) over Tint because currents Iac and Icomp are not constant. However, for the sake of clarity, the above expression is sufficient to explain the parameters which are to be taken into consideration.
This read system has certain limitations associated with the way in which the columnar compensation pattern is reproduced on the read circuit. In fact, each column has a compensation bolometer and a PMOS charge injection transistor. Imperfect reproduction of these various elements from one column to the next which is inherent in the intrinsic spatial variability of the fabrication technologies used results in non-uniform compensation efficiency. This statistical variability results in a compensation current which is not uniform from one column to the next and causes the appearance of visible columnar contrasts which thus affect the available signal.
Variation ΔRcomp on resistance Rcomp results in a current variation of the following form:
            ∂      Icomp              ∂      Rcomp        =                    Vcomp                  Rcomp          2                    ⇒              Δ        ⁢                                  ⁢        Ibolo              =                            -                      Vcomp                          Rcomp              2                                      ·        Δ            ⁢                          ⁢      Rcomp      
The compensation current can also be expressed as a function of the equation for the current in the MOSFET charge injection transistor in accordance with the following expression:
  Icomp  =                                          μ            p                    ⁢                      C            ox                          2            ·                        W          p                          L          p                      ⁢                  (                              V            SGP                    -                      V            thp                          )            2      where:                μp denotes the mobility of the positive carriers;        Cox denotes the gate oxide capacity;        Wp denotes the PMOS channel width;        Lp denotes the PMOS channel length;        VSGP denotes the gate voltage of the PMOS transistor;        VthP denotes the threshold voltage of the PMOS transistor.        
Many parameters in this equation may vary, thus producing columnar non-uniformity of the compensation current. Obviously, lithographic parameters Wp and Lp from one column to the next involve current differences. Fluctuation in the length of the channel is also one of the possible causes of this non-uniformity. There are techniques which are familiar to those skilled in the art in order to limit these variations.
In contrast, variation in threshold voltage VthP poses a problem. Assuming the threshold-voltage variation is δVth, the columnar current variation can then be expressed as follows:
      δ    ⁢                  ⁢    Icomp    =                                          μ            p                    ⁢                      C            ox                          2            ·                        W          p                          L          p                      ⁢    δ    ⁢                  ⁢                  V        thp            (                        2          ⁢                                    Icomp                                                                                          μ                      p                                        ⁢                                          C                      ox                                                        2                                ·                                                      W                    p                                                        L                    p                                                                                      +                  δ          ⁢                                          ⁢                      V            thp                              )      
Besides this static variation, the circuit shown in FIG. 1 is also sensitive to low-frequency fluctuations associated with l/f noise. The l/f noise power developed between two frequencies fmin and fmax is expressed by integrating the spectral noise density between these two bounds in accordance with the relation:
                              Icomp          2                =                              ∫                          f              ⁢                                                          ⁢              min                                      f              ⁢                                                          ⁢              max                                ⁢                                                    4                ⁢                                  k                  B                                ⁢                T                            R                        ⁢                                          R                ·                                  Vcomp                  2                                                            R                ·                f                                      ⁢                                                  ⁢                          ⅆ              f                                                              =                                            4              ⁢                              k                B                            ⁢              T                        R                    ⁢                                                    K                F                            ·              R              ·                              Vcomp                2                                      R                    ⁢                      ln            ⁡                          (                                                f                  ⁢                                                                          ⁢                  max                                                  f                  ⁢                                                                          ⁢                  min                                            )                                          where:                kB is Boltzmann's constant;        KF denotes the l/f noise coefficient of the material.        
The noise power added by the compensation structure, as expressed by the above relation, increases by a constant increment for each additional frequency decade included between the two integration bounds fmin and fmax. The compensation bolometers are permanently biased and the lower frequency bound fmin during integration can therefore be considered to be very low, to the extent that the component remains energized over an extended period once the detector is activated. Columnar interference (expressed by those skilled in the art analytically as the above noise power) manifests itself as an offset which, in the first order, is invariable from one image to the next if one considers frequencies lower than the frame frequency, but is variable over a more extended period of time if the camera in which such a detector is fitted has been operating for several minutes.
Beside this limitation, the market trend towards bolometric sensors with an increased number of pixels means that the compensation bolometers of each column, which are effective in small imagers as an absolute temperature reference, act as “local” temperature references from the point of view of the active bolometers which are the furthest away. Thus, if a thermal source of any origin whatsoever, for example circuitry elements which locally dissipate more or less heat, can influence all or some of the compensation bolometers, the latter will be influenced relatively to their distance from the source of interference, and thus reproduce a compensation current distribution which is inappropriate to the temperature variation of the substrate as seen by the sensitive bolometers which are not concerned or, generally speaking, be influenced differently by said thermal source.
The present invention relates to a detection device which uses a single compensation structure and makes it possible to overcome image quality limitations, especially those associated with differences in columnar contrast.